Space-time coding for multi-antenna ultra-wideband transmissions

ABSTRACT

Space-time (ST) coding techniques are described for multi-antenna transmissions in ultra-wideband (UWB) communication systems. The ST coding schemes may, therefore, be tailored for dense multipath channels. The techniques may be applied with linear and nonlinear modulation, coherent and noncoherent reception, and block interleaving of symbols. An UWB communication system is described that includes an ST encoder at the transmitter, multiple transmit and receive antennas, and two-step maximum ratio combining (MRC) at the receiver. The two-step MRC enables the receiver to collect full spatial and multipath diversity from a transmission. Two coding schemes for an UWB system with two transmit antennas and one receive antenna are described. Multiple antenna transmissions of ST encoded symbols increase the amount of diversity a receiver is able to collect without increasing the complexity of the receiver.

This application claims priority from U.S. Provisional Application Ser. No. 60/453,810, filed Mar. 8, 2003, the entire content of which is incorporated herein by reference.

STATEMENT REGARDING FEDERALLY SPONSORED RESEARCH OR DEVELOPMENT

This invention was made with Government support under Subcontract #497420 awarded by the University of Delaware (Army Prime #DAAD19-01-2-011). The Government may have certain rights in the invention.

TECHNICAL FIELD

The invention relates to wireless communication techniques and, in particular, techniques that employ ultra-wideband (UWB) communication.

BACKGROUND

Ultra-wideband (UWB) communication has attractive features for baseband multiple access, tactical wireless communications and multimedia services. In general, an UWB transmission consists of a train of very short pulses occupying an ultra-wide bandwidth. The information is typically encoded via either linear pulse amplitude modulation (PAM) or nonlinear pulse position modulation (PPM). The ultra-wide bandwidth includes bandwidths that are licensed from the Federal Communication Commission (FCC) for other communication purposes. However, the short pulses of the UWB transmission appear as minimal noise to non UWB-systems operating within those licensed frequencies.

Conveying information with ultra-short pulses can cause UWB transmissions to resolve many paths and become rich in multipath diversity. Consequently, rake receivers have been designed to collect the available multipath diversity to enhance the performance of UWB communication systems. Since the received UWB waveform often contains many delayed and scaled replicas of the transmitted pulses, a large number of fingers are typically needed on the Rake receiver. However, the Rake receiver may not have enough separation within the spectrum to allow each finger to track a different path of the transmission. At some point, the number of fingers on the Rake receiver may become too dense, and the receivers may become dependant on each other. In that case, no more additional diversity may be gained even if the number of fingers on the Rake receiver increases.

Moreover, each of the resolvable multipath waveforms undergoes a different channel, which causes distortion in the received pulse shapes. In some situations, the Rake receiver must know certain characteristics of each channel in order to correlate the received waveform with the delayed and scaled replicas. As a result, both the design and the implementation of Rake reception for UWB devices can be complicated. Furthermore, UWB transmissions have been shown to be very sensitive to timing jitter in non-fading channels. UWB transmissions with Rake reception are particularly sensitive to mistiming even in multipath fading channels.

SUMMARY

In general, the invention is directed to space-time (ST) coding techniques for multi-antenna transmissions in ultra-wideband (UWB) communication systems. The ST coding techniques provide effective means of enabling spatial diversity, and thus increasing channel performance and capacity within the UWB system. The UWB communication system with ST coding includes multiple transmit and receive antennas. Multiple antenna transmissions of ST-encoded symbols increase the amount of diversity a receiver is able to collect without increasing the complexity of the receiver. For example, a Rake receiver may be able to collect more diversity from a multi-antenna transmission than a single antenna transmission of the same symbol without increasing a number of fingers on the receiver.

Conventional ST coding techniques primarily focus on digital transmissions in narrowband wireless systems. In some embodiments, the invention includes analog ST coding schemes tailored for dense multipath channels. The analog coding schemes are developed for the analog UWB system to eliminate the need for sampling at the receiver. In other embodiments, the ST coding techniques may be applied with linear and nonlinear modulation, coherent and noncoherent reception, and block interleaving of symbols.

Adding one or more transmit antennas to a conventional UWB communication system with one transmit antenna and one receive antenna increases the diversity order compared to the diversity collected in the conventional UWB system. For example, an ST coding scheme that simultaneously transmits the same symbol with different waveforms from each of two transmit antennas may double the diversity the receive antenna is able to collect. In another example, an ST coding scheme that simultaneously transmits a pair of consecutive symbols with alternate orders and different waveforms from each of the two transmit antennas is able to quadruple the diversity order collected by the receive antenna.

In one embodiment, a method comprises processing a stream of information-bearing symbols to form a plurality of symbol blocks. Each symbol block comprises one or more of the information bearing symbols. The method further comprises generating multiple ultra-wideband (UWB) waveforms from the symbol blocks, wherein each of the UWB waveforms convey the symbols of their respective symbol blocks as pulses repeated over a number of frames, and transmitting the UWB waveforms over different antennas as a space-time coded UWB communication.

In another embodiment, a wireless communication device comprises a space-time (ST) encoder that processes a stream of information-bearing symbols to form a plurality of ST-encoded symbol blocks, wherein each symbol block comprises one or more of the information bearing symbols. A plurality of pulse shapers generate multiple ultra-wideband (UWB) waveforms from the symbol blocks, wherein each of the UWB waveforms convey the symbols of their respective symbol blocks as pulses repeated over a plurality of frames. A plurality of antennas transmit the UWB waveforms over a wireless communication channel.

In another embodiment, a wireless communication device comprises a plurality of antennas to receive a plurality of space-time (ST) encoded ultra wideband (UWB) waveforms through a wireless communication channel, and a maximum ratio combining (MRC) unit that processes the ST encoded UWB signals and produces a stream of estimate symbols.

In another embodiment, an ultra-wideband communication system comprises a transmitter that outputs a plurality of space-time (ST) encoded ultra wideband (UWB) waveforms via a plurality transmit antennas, and a receiver that receives the plurality of ST-encoded UWB waveforms via a wireless communication channel. The receiver performs maximum ratio combining (MRC) on the UWB signals to produce estimate symbols.

The details of one or more embodiments of the invention are set forth in the accompanying drawings and the description below. Other features, objects, and advantages of the invention will be apparent from the description and drawings, and from the claims.

BRIEF DESCRIPTION OF DRAWINGS

FIG. 1 is a block diagram illustrating an ultra-wideband (UWB) communication system.

FIG. 2 is a block diagram illustrating an example multi-antenna UWB communication system.

FIG. 3 is a block diagram illustrating an example multi-antenna UWB communication system having two transmit antennas and one receive antenna.

FIG. 4 is a flowchart illustrating an exemplary method of communication with a first space-time (ST) coding scheme (herein, “ST coding scheme I”) applied to the multi-antenna UWB communication system from FIG. 3.

FIG. 5 is a flowchart illustrating a method of communication with a second ST coding scheme (herein, “ST coding scheme II”) applied to the multi-antenna UWB communication system from FIG. 3.

FIGS. 6-14 are graphs illustrating results of simulations and comparisons that validate analyses and designs of UWB communication systems and the ST coding schemes.

DETAILED DESCRIPTION

FIG. 1 is a block diagram illustrating an ultra-wideband (UWB) communication system 2. UWB system 2 includes a transmitter 4, which communicates data to a receiver 6 by transmitting UWB waveforms through a plurality of channels 8A-8N (hereinafter, “channels 8”). Space-time (ST) coding techniques are applied in transmitter 4 to enable data symbols to be transmitted via multiple antennas. Transmitter 4 may include a plurality of transmit antennas and receiver 6 may include a plurality of receive antennas. Each of the transmit antennas corresponds to one of the channels 8 to transmit a ST-encoded UWB waveform from the transmit antenna to the receive antennas. The impulse responses of channels 8 may change from symbol to symbol.

Transmitter 4 includes an ST encoder and, in some embodiments, a frame interleaver within the ST encoder may be used to implement the ST coding schemes. The ST coding schemes may be analog and tailored for dense multipath channels. The UWB tailored ST coding schemes encode analog waveforms within symbols. ST coding for multi-antenna transmission increases performance and capacity of channels 8 by exploiting the spatial dimension, and allowing receiver 6 to collect both multipath diversity and spatial diversity.

Receiver 6 includes a maximum ratio combining (MRC) unit. The MRC unit performs MRC on UWB signals received by the receive antennas to collect spatial diversity as well as multipath diversity. In some embodiments, each of the plurality of receive antennas be associated with a corresponding Rake receiver. Each Rake receiver may generally be viewed as including L fingers configured to receive L paths of a UWB transmission waveform. In conventional UWB systems, the single antenna transmissions require a large number of fingers on the Rake receiver in order to collect enough diversity to accurately detect a received symbol. UWB system 2, as described herein, applies ST coding schemes for multi-antenna UWB transmissions to increase an amount of diversity collected at the receiver without increasing the overall required number of Rake fingers.

Transmitter 4 processes a stream of information-bearing symbols and transmits each ST-encoded data symbol as a train of very short pulses to receiver 6 using a modulation format, such as pulse amplitude modulation (PAM) or pulse position modulation (PPM). PAM is a linear modulation technique that requires knowledge of channels 8 at receiver 6. PPM is a non-linear modulation technique that does not generally require knowledge of channels 8 at receiver 6 when orthogonal PPM, or on-off keying (OOK), is employed.

FIG. 2 is a block diagram illustrating in further detail the example multi-antenna UWB communication system 2 (FIG. 1) in which transmitter 4 transmits ST-encoded UWB waveforms to receiver 6 via channels 8. In the illustrated embodiment, transmitter 4 comprises a ST encoder 12 that produces a stream of ST-encoded data symbols, and two or more transmit antennas. For each antenna, transmitter 4 includes a power loader 14 to power load the respective encoded stream of symbols produced by ST encoder 12, and a pulse shaper 16 that generates UWB pulse trains for transmitting the ST-encoded UWB signal through the respective one of the transmit antennas. An overall channel effect (herein, “the overall channel”) may be represented as the convolution of the pulse shaper 16 and a respective physical multipath channel 17.

Receiver 6 comprises one or more Rake receivers 20, a MRC unit 22, and a symbol detector 24. The number of Rake receivers does not necessarily equal the number of transmit antennas included in transmitter 4. Rake receivers 20 receive the transmitted symbol as a noisy waveform, and utilize an input pulse waveform 19 for correlation with the received waveform. MRC unit 22 accepts the output of the one or more Rake receivers 20 and performs MRC to yield a decision statistic. Symbol detector 24 uses the decision statistic to determine an estimate of the original transmitted data symbols.

The UWB system 2, including the modulation, channel model, receiver structure, and detection method, will be described herein through the analysis of a single antenna transmission using PAM. The performance of single antenna transmissions not only serves as motivation to include ST coding for UWB multi-antenna communications, but also provides a benchmark for multi-antenna performance comparisons.

In general, transmitter 4 conveys the stream of binary information symbols as a stream of ultra-short pulses. With N_(t) denoting the number of transmit antennas, every binary symbol s=±1 is power loaded by power loader 14, pulse shaped by pulse shaper 16, and transmitted repeatedly over N_(f) consecutive frames, each of duration T_(f). Pulse shaper 16 employs a pulse waveform w(t) with a typical duration T_(w) between 0.2 ns to 2 ns, which results in a transmission occupying an ultra-wide bandwidth. The physical multipath channel g(t) 17 can be expressed in terms of multipath delays and gains as:

$\begin{matrix} {{g(t)} = {\sum\limits_{l_{g} = 0}^{L_{g} - 1}{{\alpha_{g}\left( l_{g} \right)}{\delta\left( {t - {\tau_{g}\left( l_{g} \right)}} \right)}}}} & (1) \end{matrix}$ where L_(g) is a number of transmission paths and τ_(g)(l_(g))>τ_(g)(l_(g)−1), ∀l_(g)∈[1L _(g)−1]. The overall channel h(t) 18 comprises the convolution of the pulse waveform w(t) of pulse shaper 16 with the physical multipath channel g(t) 17, and is given by:

$\begin{matrix} {{h(t)}:={{{g(t)}*{w(t)}} = {\sum\limits_{l_{g} = 0}^{L_{g} - 1}{{\alpha_{g}\left( l_{g} \right)}{w\left( {t - {\tau_{g}\left( l_{g} \right)}} \right)}}}}} & (2) \end{matrix}$ where * stands for convolution. With T_(g):=τ_(g)(L_(g)−1) denoting the maximum delay spread of the dense multipath channel, we avoid intersymbol interferences (ISI) by simply choosing T_(f)≧T_(g)+T_(w).

The Rake receiver 20 has L fingers, and employs pulse waveform w(t) 19, which is substantially similar to the pulse waveform used in pulse shaper 16, as the correlator template. MRC is performed by MRC unit 22 at the receiver 6 to yield a decision statistic. Based on the decision statistic, an estimate of the transmitted symbol ŝ is formed by the detector 24. The multipath fading channel 17 is modeled as quasi-static, which is typical for an indoor environment. More precisely, the overall channel h(t) 18 is assumed to remain invariant over a symbol duration N_(f)T_(f) seconds, but it is allowed to change from symbol to symbol.

When a single transmit antenna is deployed, the binary symbol s is transmitted with power ε, using the waveform

$\begin{matrix} {{{s(t)} = {s\sqrt{\frac{ɛ}{N_{f}}}{\sum\limits_{n_{f} = 0}^{N_{f} - 1}{w\left( {t - {n_{f}T_{f}}} \right)}}}},{s = {\pm 1}}} & (3) \end{matrix}$ where the pulse waveform w(t) has unit energy, i.e., ∫₀ ^(T) ^(f) w²(t)dt=1. With a single receive antenna, and assuming that timing offsets have been compensated accurately, the received noisy waveform corresponding to transmitted symbol s is given by:

$\begin{matrix} {{r(t)} = {{{{s(t)}*{g(t)}} + {\eta(t)}} = {{s\sqrt{\frac{ɛ}{N_{f}}}{\sum\limits_{n_{f} = 0}^{N_{f} - 1}{h\left( {t - {n_{f}T_{f}}} \right)}}} + {\eta(t)}}}} & (4) \end{matrix}$ where η(t) is the additive white Gaussian noise (AWGN) with zero mean and variance σ².

The received UWB waveform contains a large number of resolvable multipath components, L_(g), due to the ultra-short duration of the pulse waveform w(t). In order to harvest the multipath diversity, Rake receiver 20 is employed at the receiver 6. Using the pulse waveform w(t) 19 as a reference, Rake receiver 20 with L fingers yields the correlation of the received waveform r(t) with L delayed versions of the reference waveform, namely {w(t−τ(l))}_(t=0) ^(L−1), where 0≦τ(0)<τ(1)< . . . <τ(L−1)≦T_(g). Notice that {T_(g) (l_(g))}l_(g)=0 L_(g)−1 in equation (2) denotes the arrival times of the physical multipath components, which are merely determined by the physical environment. Therefore, no restrictions apply to the number and/or intervals of τ_(g)(l_(g)). On the other hand, the matched filter employing the reference w(t) can not resolve multipath components whose delays differ less than one pulse duration T_(w). The arrival times of the physical multipath components are selected such that τ(l)−τ(l−1)≧2T_(w),∀l∈[0,L−1]. The L fingers of Rake receiver 20 are selected such that τ(l)=2lT_(w),∀l∈[0,L−1]. Therefore, L≦L_(g) must hold true.

During each frame duration T_(f), the output of the l-th finger of the Rake receiver 20 is given by:

$\begin{matrix} {{{x(l)} = {{s\sqrt{\frac{ɛ}{N_{f}}}{\alpha(l)}} + {\zeta(l)}}},{\forall{l \in \left\lbrack {0,{L - 1}} \right\rbrack}}} & (5) \end{matrix}$ where ζ(l):=∫₀ ^(T) ^(f) η(t)w(t−τ(l))dt, and

$\begin{matrix} {{\alpha(l)}:={{\int_{0}^{T_{f}}{{h(t)}{w\left( {t - {\tau(l)}} \right)}{\mathbb{d}t}}} = {\sum\limits_{l_{g} = 0}^{L_{g} - 1}{{\alpha_{g}\left( l_{g} \right)}{R_{w}\left( {{\tau(l)} - {\tau_{g}\left( l_{g} \right)}} \right)}}}}} & (6) \end{matrix}$ with R_(w)(τ)=∫₀ ^(T) ^(f) w(t)w(t−τ)dt denoting the autocorrelation function of w(t). It is evident that ζ(l) has zero mean and variance σ², since w(t) has unit energy. Also recall that the finger delays satisfy τ(l)−τ(l−1)≧T_(w),∀l∈[l,L−1]; hence, ζ(l) is also white.

To maximize the signal-to-noise ratio (SNR), MRC unit 22 is used to collect the spatial diversity. In this case, MRC is employed in two levels: i) the MRC of L fingers the Rake receiver 20 per frame; and ii) the MRC of the N_(f) frames corresponding to the same symbol. To apply MRC, the receiver 6 requires knowledge of {α(l)}_(l=0) ^(L−1). Recalling the expression in equation (6), the receiver 6 requires both the multipath delays and gains. In other words, the physical multipath channel g(t) 17 needs to be acquired through, e.g., the transmission of pilot waveforms. Assuming that the receiver 6 has perfect knowledge of {α(l)}_(l=0) ^(L−1), the output of MRC unit 22 per received frame n_(f)∈[1,N_(f)] is:

$\begin{matrix} \begin{matrix} {{y\left( n_{f} \right)} = {\sum\limits_{l = 0}^{L - 1}{{x(l)}{\alpha(l)}}}} \\ {= {{s\sqrt{\frac{ɛ}{N_{f}}}{\sum\limits_{l = 0}^{L - 1}{\alpha^{2}(l)}}} + {\sum\limits_{l = 0}^{L - 1}{{\alpha(l)}{\zeta(l)}}}}} \\ {{= {s\sqrt{{\frac{ɛ}{N_{f}}ɛ_{g}} + {\xi\left( n_{f} \right)}}}},{\forall{n_{f} \in \left\lbrack {0,{N_{f} - 1}} \right\rbrack}}} \end{matrix} & (7) \end{matrix}$ where ε_(g):=Σ_(l=0) ^(L−1)α²(l), and ξ(n_(f)):=Σ_(l=0) ^(L−1)α(l)ζ(l). Notice that ε_(g) represents the energy captured by the L fingers of Rake receiver 20. For fixed L, ε_(g) is determined by the multipath channel g(t) 17, since pulse waveform w(t) was designed to have unit energy. Also notice that ξ(n_(f)) is still a white Gaussian noise with zero mean, but its variance is now given by ε_(g)σ².

With the overall channel 18 remaining invariant over a symbol duration N_(f)T_(f), the MRC of N_(f) frames amounts to summing up {y(n_(f))}_(n) _(f) ₌₀ ^(N) ⁻¹ from equation (7). The resulting decision statistic corresponding to the symbol s is given below.

$\begin{matrix} {z = {{s\sqrt{N_{f}ɛ}ɛ_{g}} + {\sum\limits_{n_{f} = 0}^{N_{f} - 1}{\xi\left( n_{f} \right)}}}} & (8) \end{matrix}$ The white Gaussian noise in equation (8) has zero mean, and variance N_(f)ε_(g)σ². When the maximum likelihood (ML) detector is used, a bit error ratio (BER) is given by: P(error|{α(l)}_(l=0) ^(L−1))=Q(√{square root over (ρε_(g))})  (9) where ρ:=ε/σ² denotes the transmitted SNR, and Q(x):=(1/√{square root over (2π)})∫_(x) ^(∞) exp( −t²/2)dt is the Gaussian tail function. Conditioned on ε_(g), the Chernoff bound yields: P(error|{α(l)}_(l=0) ^(L−1))≦exp(−ρε_(g)/2)  (10) or, using the definition of ε_(g) in equation (7), P(error|{α(l)}_(l=0) ^(L−1))≦exp(−ρΣ_(l=0) ^(L−1)α²(l)/2)=π_(l=0) ^(L−1) exp(−ρα²(l)/2).  (11)

In indoor environments with multiple reflections and refractions, the gain of each path L_(g) can be modeled as a Rayleigh distributed random variable, while the phase is a uniformly distributed random variable. Since UWB systems employ real signals, only the real part of each path gain is of interest, which has Gaussian distribution with zero mean. As combinations of Gaussian random variables, α(l)'s are also Gaussian distributed. If the finger delays are chosen such that τ(l)−τ(l−1)≧2T_(w), ∀l∈[1,L−1], then E[α(l₁)α(l₂)]=0, ∀l₁≠l₂. In other words, α(l₁) and α(l₂) are uncorrelated ∀l₁≠l₂∈[0,L−1]. Letting β(l):=E[α²(l)], averaging the conditional BER over the independent Gaussian distributions of α(l) yields the average BER bounded as shown below.

$\begin{matrix} {{{P({error})} \leq {\prod\limits_{l = 0}^{L - 1}{E\left\lbrack {\exp\left( {{- \rho}\;{{\alpha^{2}(l)}/2}} \right)} \right\rbrack}}} = {\prod\limits_{l = 0}^{L - 1}\left( {1 + {\rho\;{\beta(l)}}} \right)^{- \frac{1}{2}}}} & (12) \end{matrix}$ At high SNR (ρ>>), the upper bound is given by:

$\begin{matrix} {{{P({error})} \leq \left( {\rho^{L}{\prod\limits_{l = 0}^{L - 1}{\beta(l)}}} \right)^{- \frac{1}{2}}} = \left( {\beta_{L}\rho} \right)^{- \frac{L}{2}}} & (13) \end{matrix}$ where coding gain β_(L):=(π_(l=0) ^(L−1)β(l))^(1/L).

It may also be verified the the BER upper bound in equation (13) becomes

$\left( \frac{\rho\;\beta_{L}}{2} \right)^{- L}$ if α(l)'s are independent complex Gaussian random variables with variance β(l)/2. Equation (13) confirms that as the number of fingers L increases, the diversity order also increases. The increase in L can come from either denser finger delays, or larger finger delays. With dense τ(l)'s, the mutual independence among α(l)'s becomes invalid. With larger τ(l), the generally decreasing power profile of the multipath channel 17 will decrease the coding gain β_(L). In fact, the diversity order comes from the energy capture of the Rake receiver 20. The energy capture however, does not increase linearly with the number of fingers L. As a result, large L does not benefit performance, but increases the implementation complexity at the receiver. Therefore, a large number of fingers L is formidable, while performance requirements are yearning for higher diversity order.

FIG. 3 is a block diagram illustrating an example multi-antenna UWB communication system 30 that has two transmit antennas 36A, 36B and a rake receiver 40. FIG. 3 illustrates a specific example of the UWB system 2 more generally illustrated in FIGS. 1 and 2. In system 30, transmitter 4 ST-encodes data 32 and transmits ST-encoded UWB waveforms via channels 8 to receiver 6, which outputs estimated data 45. The ST coding schemes may be analog for use with the analog UWB system 30 to eliminate the need for sampling at the receiver 6.

Transmitter 4 includes an ST encoder 34, a first transmit antenna 36A, and a second transmit antenna 36B. Substantially similar to the single antenna transmission described above, every binary symbol s=±1 of data 32 is power loaded and pulse shaped before being transmitted repeatedly over N_(f) consecutive frames of duration T_(f). Channels 8 include a first multipath channel 38A, which carries the transmission signal waveform from first transmit antenna 36A, and a second multipath channel 38B, which carries the transmission signal waveform from second transmit antenna 36B. Receiver 6 includes a Rake receiver 40 with L fingers, MRC unit 42, and a detector 44, which generates estimated data 45.

A plurality of ST coding schemes may be applied to system 30 to encode analog transmission waveforms within data symbols to improve the diversity order compared to the single antenna transmission case described above. For example, one ST coding scheme may duplicate the symbols and use first and second transmit antennas 36 to simultaneously transmit the same symbol. Another ST coding scheme may duplicate pairs of consecutive symbols and use first and second transmit antennas 36 to simultaneously transmit the symbol pairs with alternate orders. In either case, transmission power and transmission rate per symbol may remain the same as in the single antenna transmission. In other embodiments, a multi-antenna UWB communication system may comprise any number of transmit and receive antennas and may require ST coding schemes that accommodate the number of antennas within the system.

FIG. 4 is a flowchart illustrating a method of communication with an exemplary first coding scheme (ST coding scheme I) applied to the multi-antenna UWB communication system 30 from FIG. 3. ST coding scheme I transmits the same symbol over both transmit antennas 36A, 36B. In particular, ST encoder 34 duplicates each symbol in a stream of information-bearing symbols to form a first symbol block and a second symbol block (step 50). Each symbol block is then power loaded with power P/2 to ensure that a total transmit power for the symbol is equal to the single antenna transmission. Each of the symbol blocks is pulse shaped with pulse waveform w(t) to generate a first UWB waveform for the first symbol block and a second UWB waveform for the second symbol block (step 51). During each symbol duration N_(f)T_(f), transmitter 4 simultaneously transmits the first symbol block waveform

$\begin{matrix} {{s_{0}(t)} = {s\sqrt{\frac{ɛ}{2N_{f}}}{\sum\limits_{n_{f} = 0}^{N_{f} - 1}\;{\left( {- 1} \right)^{n_{f}}{w\left( {t - {n_{f}T_{f}}} \right)}}}}} & (14) \end{matrix}$ from the first transmit antenna 36A through channel g₀(t) 38A, and transmits the second symbol block waveform

$\begin{matrix} {{s_{1}(t)} = {s\sqrt{\frac{ɛ}{2N_{f}}}{\sum\limits_{n_{f} = 0}^{N_{f} - 1}{w\left( {t - {n_{f}T_{f}}} \right)}}}} & (15) \end{matrix}$ from the second transmit antenna 36B through channel g₁(t) 38B (step 52). In that way, the symbol is transmitted over N_(f) frames.

During the symbol duration, Rake receiver 40 receives a noisy waveform of the transmitted symbol block waveforms (step 54). The received waveform is given below.

$\begin{matrix} \begin{matrix} {{r(t)} = {{{s_{0}(t)}*{g_{0}(t)}} + {{s_{1}(t)}*{g_{1}(t)}} + {\eta(t)}}} \\ {= {{s\sqrt{\frac{ɛ}{2N_{f}}}{\sum\limits_{n_{f} = 0}^{N_{f} - 1}\left\lbrack {{\left( {- 1} \right)^{n_{f}}{h_{0}\left( {t - {n_{f}T_{f}}} \right)}} + {h_{1}\left( {t - {n_{f}T_{f}}} \right)}} \right\rbrack}} + {\eta(t)}}} \end{matrix} & (16) \end{matrix}$ The received waveform may be separated into even and odd indexed frames of the symbol s as r_(e)(t) and r_(o)(t), respectively, to reduce a complexity of Rake receiver 40.

$\begin{matrix} {{{r(t)} = {\sum\limits_{n_{f} = 0}^{N_{f}^{\prime} - 1}\;\left\lbrack {{r_{e}\left( {t - {2n_{f}T_{f}}} \right)} + {r_{o}\left( {t - {2n_{f}T_{f}} - T_{f}} \right)}} \right\rbrack}},{{{where}\mspace{14mu} N_{f}^{\prime}} = \frac{N_{f}}{2}}} & \left( {17a} \right) \\ {{{r_{e}(t)} = {{s{\sqrt{\frac{ɛ}{2N_{f}}}\left\lbrack {{h_{0}(t)} + {h_{1}(t)}} \right\rbrack}} + {{\eta_{e}(t)}\mspace{14mu}{and}}}}\text{}{{r_{o}(t)} = {{s{\sqrt{\frac{ɛ}{2N_{f}}}\left\lbrack {{h_{1}(t)} + {h_{0}(t)}} \right\rbrack}} + {\eta_{o}(t)}}}} & \left( {17b} \right) \end{matrix}$

Inputting the even and odd indexed frames of the symbol into Rake receiver 40, the output of the l-th finger is given by:

$\begin{matrix} {{x_{e}(l)} = {{s{\sqrt{\frac{ɛ}{2N_{f}}}\left\lbrack {{\alpha_{0}(l)} + {\alpha_{1}(l)}} \right\rbrack}} + {{\zeta_{e}(l)}\mspace{14mu}{for}\mspace{14mu}{even}\mspace{14mu}{frames}}}} & (18) \\ {{x_{o}(l)} = {{s{\sqrt{\frac{ɛ}{2N_{f}}}\left\lbrack {{\alpha_{1}(l)} - {\alpha_{0}(l)}} \right\rbrack}} + {{\zeta_{o}(l)}\mspace{14mu}{for}\mspace{14mu}{odd}\mspace{14mu}{frames}}}} & \; \end{matrix}$ where

α_(m)(l) := ∫₀^(T_(f))h_(m)(t)ω(t − τ(l)) 𝕕t  for  m = 0, 1. MRC unit 42 accepts the output of the L fingers of Rake receiver 40. MRC is performed on each frame of the ST-encoded signal with the output given below.

$\begin{matrix} {{y_{e}\left( n_{f} \right)} = {{s\sqrt{\frac{ɛ}{2N_{f}}}{\sum\limits_{l = 0}^{L - 1}\;\left\lbrack {{\alpha_{0}(l)} + {\alpha_{1}(l)}} \right\rbrack^{2}}} + {{\xi_{e}\left( n_{f} \right)}\mspace{14mu}{for}\mspace{14mu}{even}\mspace{14mu}{frames}}}} & (19) \\ {{y_{o}\left( n_{f} \right)} = {{s\sqrt{\frac{ɛ}{2N_{f}}}{\sum\limits_{l = 0}^{L - 1}\;\left\lbrack {{\alpha_{1}(l)} + {\alpha_{0}(l)}} \right\rbrack^{2}}} + {{\xi_{o}\left( n_{f} \right)}\mspace{14mu}{for}\mspace{14mu}{odd}\mspace{14mu}{frames}}}} & \; \end{matrix}$ Notice that

${{\xi_{e}\left( n_{f} \right)}:={\sum\limits_{l = 0}^{L - 1}\;{\left\lbrack {{\alpha_{0}(l)} + {\alpha_{1}(l)}} \right\rbrack{\zeta_{e}(l)}}}},{and}$ ${\xi_{o}\left( n_{f} \right)}:={\sum\limits_{l = 0}^{L - 1}\;{\left\lbrack {{\alpha_{1}(l)} - {\alpha_{0}(l)}} \right\rbrack{\zeta_{o}(l)}}}$ are white Gaussian noise variables with zero mean and variances

$\sigma_{\xi_{e}}^{2} = {\sigma^{2}{\sum\limits_{l = 0}^{L - 1}\;\left\lbrack {{\alpha_{0}(l)} + {\alpha_{1}(l)}} \right\rbrack^{2}}}$ and

${\sigma_{\xi_{o}}^{2} = {\sigma^{2}{\sum\limits_{l = 0}^{L - 1}\left\lbrack {{\alpha_{1}(l)} - {\alpha_{0}(l)}} \right\rbrack^{2}}}},$ respectively, ∀n_(f)∈[0,N_(f)′−1].

MRC is then performed on all the frames of the ST-encoded signal combined by summing y_(e)(n_(f)) and y_(o)(n_(f)) over the N_(f) frames corresponding to the symbol s to yield a decision statistic (step 56)

$\begin{matrix} {z = {{s\sqrt{\frac{N_{f}ɛ}{2}}\left( {ɛ_{g0} + ɛ_{g1}} \right)} + {\sum\limits_{n_{f} = 0}^{N_{f}^{\prime} - 1}\left( {{\xi_{e}\left( n_{f} \right)} + {\xi_{o}\left( n_{f} \right)}} \right)}}} & (20) \end{matrix}$ where

${ɛ_{gm} = {\sum\limits_{l = 0}^{L - 1}{\alpha_{m}^{2}(l)}}},$ and the zero-mean noise has variance given by N_(f)σ²(ε_(g0)+ε_(g1)). Detector 44 then estimates symbol s based on the decision statistic (step 58). Averaging over {α₀(l),α₁(l)}_(l=0) ^(L=1), the average bit error rate (BER) is bounded by

$\begin{matrix} {{P({error})} \leq \left( {\frac{\beta_{L}}{2}\rho} \right)^{- L}} & (21) \end{matrix}$ at high SNR. Compared to equation (13) of the single antenna transmission case, ST coding scheme I doubles the diversity order while losing 3 dB coding gain by employing N_(t)=2 transmit antennas.

FIG. 5 is a flowchart illustrating another exemplary method of communication with ST coding (ST coding scheme II) applied to the multi-antenna UWB communication system 30 from FIG. 3. In general, ST coding scheme II transmits a block of two symbols s_(a) and s_(b) over transmit antennas 36A, 36B in alternate order. More specifically, ST encoder 34 encodes the symbol pair into a first symbol block and a second symbol block with alternate symbol orders in each symbol block (step 60). Each of the symbol blocks is power loaded to ensure that a transmit power of each symbol is equal to the single antenna transmission power for each symbol. Each of the symbol blocks is pulse shaped with pulse waveform w(t) to generate a first UWB waveform for the first symbol block and a second UWB waveform for the second symbol block (step 61). Over two symbol durations 2N_(f)T_(f), transmitter 4 transmits the first symbol block waveform

$\begin{matrix} {{s_{0}(t)} = {s\sqrt{\frac{ɛ}{2N_{f}}}{\sum\limits_{n_{f} = 0}^{N_{f} - 1}\left\lbrack {{s_{a}{w\left( {t - {2n_{f}T_{f}}} \right)}} - {s_{b}{w\left( {t - {2n_{f}T_{f}} - T_{f}} \right)}}} \right\rbrack}}} & (22) \end{matrix}$ from the first transmit antenna 36A through channel 38A, and transmits the second symbol block waveform

$\begin{matrix} {{s_{1}(t)} = {s\sqrt{\frac{ɛ}{2N_{f}}}{\sum\limits_{n_{f} = 0}^{N_{f} - 1}\left\lbrack {{s_{b}{w\left( {t - {2n_{f}T_{f}}} \right)}} - {s_{a}{w\left( {t - {2n_{f}T_{f}} - T_{f}} \right)}}} \right\rbrack}}} & (23) \end{matrix}$ from the second transmit antenna 36B through channel 38B (step 62). In that way, each symbol is transmitted over N_(f) frames.

During a first symbol duration, Rake receiver 40 receives a first waveform of the transmit signals (step 64)

$\begin{matrix} {{r(t)} = {{\sqrt{\frac{ɛ}{2{Nf}}}{\sum\limits_{n_{f} = 0}^{N_{f}^{\prime} - 1}\left\lbrack \begin{matrix} {{s_{a}{h_{0}\left( {t - {2n_{f}T_{f}}} \right)}} +} \\ {{s_{b}{h_{1}\left( {t - {2n_{f}T_{f}}} \right)}} +} \\ {{s_{a}{h_{1}\left( {t - {2n_{f}T_{f}} - T_{f}} \right)}} -} \\ {s_{b}{h_{0}\left( {t - {2n_{f}T_{f}} - T_{f}} \right)}} \end{matrix} \right\rbrack}} + {\eta(t)}}} & (24) \end{matrix}$ where g₀(t) and g₁(t) denote the impulse responses from the first and second transmit antennas 36, respectively, to the Rake receiver 40 during the first symbol duration and h₀(t) and h_(l)(t) denote the corresponding overall channels. During a second symbol duration, Rake receiver 40 receives a second waveform of the transmit signals (step 66)

$\begin{matrix} {{r^{\prime}(t)} = {{\sqrt{\frac{P}{2{Nf}}}{\sum\limits_{n_{f} = 0}^{N_{f}^{\prime} - 1}\begin{bmatrix} {{s_{a}{h_{0}^{\prime}\left( {t - {2n_{f}T_{f}}} \right)}} +} \\ {{s_{b}{h_{1}^{\prime}\left( {t - {2n_{f}T_{f}}} \right)}} +} \\ {{s_{a}{h_{1}^{\prime}\left( {t - {2n_{f}T_{f}} - T_{f}} \right)}} -} \\ {s_{b}{h_{0}^{\prime}\left( {t - {2n_{f}T_{f}} - T_{f}} \right)}} \end{bmatrix}}} + {\eta(t)}}} & (25) \end{matrix}$ where g₀′(t) and g_(l)′(t) denote the impulse responses from the first and second transmit antennas 36, respectively, to the Rake receiver 40 during the second symbol duration and h₀′(t) and h_(l)′(t) denote the corresponding overall channels.

The first received waveform, equation (24), may be separated into even and odd indexed frames as r_(e)(t) and r_(o)(t), respectively, to reduce complexity of Rake receiver 40. The waveform is given below.

$\begin{matrix} {{r(t)} = {\sum\limits_{n_{f} = 0}^{N_{f}^{\prime} - 1}\left\lbrack {{r_{e}\left( {t - {2n_{f}T_{f}}} \right)} + {r_{o}\left( {t - {2n_{f}T_{f}} - T_{f}} \right)}} \right\rbrack}} & (26) \\ {{r_{e}(t)} = {{\sqrt{\frac{ɛ}{2N_{f}}}\left\lbrack {{s_{a}{h_{0}(t)}} + {s_{b}{h_{1}(t)}}} \right\rbrack} + {{\eta_{e}(t)}\mspace{14mu}{and}}}} & (27) \\ {{r_{o}(t)} = {{\sqrt{\frac{ɛ}{2N_{f}}}\left\lbrack {{s_{a}{h_{1}(t)}} - {s_{b}{h_{0}(t)}}} \right\rbrack} + {\eta_{o}(t)}}} & \; \end{matrix}$ Inputting the even and odd indexed frames of the first waveform into Rake receiver 40, the output of the l-th finger is given below.

$\begin{matrix} \begin{matrix} {{x_{e}(l)} = {{\sqrt{\frac{ɛ}{2N_{f}}}\left\lbrack {{s_{a}{\alpha_{0}(l)}} + {s_{b}{\alpha_{1}(l)}}} \right\rbrack} + {{\zeta_{e}(l)}\mspace{14mu}{for}\mspace{14mu}{even}\mspace{14mu}{frames}}}} \\ {{x_{o}(l)} = {{\sqrt{\frac{ɛ}{2N_{f}}}\left\lbrack {{s_{a}{\alpha_{1}(l)}} - {s_{b}{\alpha_{0}(l)}}} \right\rbrack} + {{\zeta_{o}(l)}\mspace{14mu}{for}\mspace{14mu}{even}\mspace{14mu}{frames}}}} \end{matrix} & (28) \end{matrix}$ MRC unit 42 accepts the output of the L fingers of Rake receiver 40 for the first waveform. MRC is performed on each frame of each ST-encoded signal with the output given below.

$\begin{matrix} \begin{matrix} {{y_{a}\left( n_{f} \right)} = {\sum\limits_{l = 0}^{L - 1}\left\lbrack {{{\alpha_{0}(l)}{x_{e}(l)}} + {{\alpha_{1}(l)}{x_{o}(l)}}} \right\rbrack^{2}}} \\ {= {{s_{a}\sqrt{\frac{ɛ}{2N_{f}}}\left( {ɛ_{g0} + ɛ_{g1}} \right)} + {\xi_{a}\left( n_{f} \right)}}} \\ {{y_{b}\left( n_{f} \right)} = {\sum\limits_{l = 0}^{L - 1}\left\lbrack {{{\alpha_{1}(l)}{x_{e}(l)}} - {{\alpha_{0}(l)}{x_{o}(l)}}} \right\rbrack^{2}}} \\ {= {{s_{b}\sqrt{\frac{ɛ}{2N_{f}}}\left( {ɛ_{g0} + ɛ_{g1}} \right)} + {\xi_{b}\left( n_{f} \right)}}} \end{matrix} & (29) \end{matrix}$ Notice that

$\begin{matrix} {{{\xi_{a}\left( n_{f} \right)}:={\sum\limits_{l = 0}^{L - 1}\left\lbrack {{{\alpha_{0}(l)}{\zeta_{e}(l)}} + {{\alpha_{1}(l)}{\zeta_{o}(l)}}} \right\rbrack}},{and}} \\ {{\xi_{b}\left( n_{f} \right)}:={\sum\limits_{l = 0}^{L - 1}\left\lbrack {{{\alpha_{1}(l)}{\zeta_{e}(l)}} - {{\alpha_{0}(l)}{\zeta_{o}(l)}}} \right\rbrack}} \end{matrix}$ are white Gaussian noise variables with zero mean and variances σ_(ξa) ²=σ_(ξb) ²=σ²(ε_(g0)+ε_(g1)).

MRC is then performed on all the frames of each ST-encoded signal for the first waveform combined by summing equations (29) over the first N_(f) frames to yield a first decision statistic for each symbol (step 68)

$\begin{matrix} \begin{matrix} {z_{a} = {{s_{a}\sqrt{\frac{N_{f}ɛ}{8}}\left( {ɛ_{g0} + ɛ_{g1}} \right)} + {\sum\limits_{n_{f} = 0}^{N_{f}^{\prime} - 1}{\xi_{a}\left( n_{f} \right)}}}} \\ {z_{b} = {{s_{b}\sqrt{\frac{N_{f}ɛ}{8}}\left( {ɛ_{g0} + ɛ_{g1}} \right)} + {\sum\limits_{n_{f} = 0}^{N_{f}^{\prime} - 1}{\xi_{b}\left( n_{f} \right)}}}} \end{matrix} & (30) \end{matrix}$ where the two noise terms have identical variance N_(f)σ²(ε_(g0)+ε_(g1))/2. As shown above, MRC unit 42 separates the outputs corresponding to the two symbols and decouples the detection of s_(a) and s_(b). After carrying out the same steps on the second received waveform, equation (25), MRC unit 42 yields a second decision statistic for each symbol (step 70)

$\begin{matrix} \begin{matrix} {z_{a}^{\prime} = {{s_{a}\sqrt{\frac{N_{f}ɛ}{8}}\left( {ɛ_{{g0}^{\prime}} + ɛ_{{g1}^{\prime}}} \right)} + {\sum\limits_{n_{f} = 0}^{N_{f}^{\prime} - 1}{\xi_{a}^{\prime}\left( n_{f} \right)}}}} \\ {z_{b}^{\prime} = {{s_{b}\sqrt{\frac{N_{f}ɛ}{8}}\left( {ɛ_{{g0}^{\prime}} + ɛ_{{g1}^{\prime}}} \right)} + {\sum\limits_{n_{f} = 0}^{N_{f}^{\prime} - 1}{\xi_{b}^{\prime}\left( n_{f} \right)}}}} \end{matrix} & (31) \end{matrix}$ where the variance of the two noise terms is N_(f)σ²(ε_(g0′)+ε_(g1′))/2.

The first and second decision statistics, z_(a) and z_(a)′, for symbol s_(a) are combined and the first and second decision statistics, z_(b), and z_(b)′, from symbol s_(b) are also combined (step 72). Detector 44 then estimates symbols s_(a) and s_(b) based on the decision statistics (step 74). Averaging over {α₀(l),α₁(l),α₀′(l),α₁′(l)}_(l=0) ^(L−1), the average BER is bounded by

$\begin{matrix} {{P({error})} \leq \left( {\frac{\beta_{L}}{4}\rho} \right)^{2L}} & (32) \end{matrix}$ at high SNR. Compared to equation (21) of ST coding scheme I, ST coding scheme II provides twice the diversity order without increasing the number of Rake receiver fingers or the channel estimation burden. However, ST coding scheme II also loses 3 dB coding gain.

The two analog ST coding schemes were described above as being applied to PAM. In other embodiments, the coding schemes may be applied to binary PPM. With binary PPM, a symbol −1 is represented by the pulse waveform w(t) and a symbol +1 is represented by a delayed pulse waveform w(t−Δ). To avoid ISI, the frame duration is chosen such that T_(f)≧T_(g)+T_(w)+Δ. The delay Δ may be chosen to minimize the correlation

∫₀^(T_(f))w(t)w(t − Δ)𝕕t, which yields Δ=0.156 ns. The delay may also be chosen to create an orthogonal PPM by setting the same correlation function to zero. Any delay greater than or equal to T_(w) will result in orthogonal PPM, but choosing Δ=T_(g)+T_(w) results in an on-off keying (OOK). OOK ensures the orthogonality of the modulation even after propagation through frequency-selective channels with maximum delay spread up to T_(g). However, with the same pulse amplitude and symbol SNR, OOK results in approximately half the transmission rate of PAM or PPM with small Δ.

ST coding scheme I described in reference to FIG. 4, can be applied to PPM with an arbitrary delay without modification. However, ST coding scheme II described in reference to FIG. 5, can only be applied to OOK signaling. When OOK signaling is not applied, multipath propagation destroys orthogonality between the pulse waveform and the delayed pulse waveform, which prevents decoupling of s_(a) and s_(b).

When OOK is applied to the ST coding schemes, noncoherent reception becomes possible as the diversity collection and symbol detection can be performed without channel information. The orthogonal, nonlinear PPM is exploited to guarantee symbol detectability and enable full spatial diversity. After modification to accommodate PPM, essentially the same steps as laid out above for PAM may be taken to yield a decision statistic. For example, the modified transmit signals for ST coding scheme I for OOK are given by:

${s_{0}(t)} = {\sqrt{\frac{ɛ}{2N_{f}}}{\sum\limits_{n_{f} = 0}^{N_{f} - 1}{\left( {- 1} \right)^{n_{f}}{w\left( {t - {n_{f}T_{f}} - {\overset{\sim}{s}\;\Delta}} \right)}\mspace{14mu}{and}}}}$ ${s_{1}(t)} = {\sqrt{\frac{ɛ}{2N_{f}}}{\sum\limits_{n_{f} = 0}^{N_{f} - 1}{w\left( {t - {n_{f}T_{f}} - {\overset{\sim}{s}\;\Delta}} \right)}}}$ where {tilde over (s)}:=(s+1)/2. Energy at each symbol duration may be determined such that the highest energy decision statistic is used to determine the estimated symbol ŝ=arg max_(s) ₀ z(s₀), or in the case of ST coding scheme II (ŝ_(a),ŝ_(b))=arg max_((s) ₁ _(,s) ₂₎ z(s₁,s₂).

In some embodiments, the ST coding schemes may be implemented differently than presented above. For example, antenna switching may be used to transmit the encoded symbols from the transmitter. Antenna switching allows one transmit antenna to transmit with full power while the other transmit antenna is shut off, and vise versa. For example, the transmit signals of ST coding scheme I for this case are given by

$\begin{matrix} {{s_{0}(t)} = {s\sqrt{\frac{ɛ}{N_{f}}}{\sum\limits_{n_{f} = 0}^{N_{f}^{\prime} - 1}{{w\left( {t - {2n_{f}T_{f}}} \right)}\mspace{14mu}{and}}}}} \\ {{s_{1}(t)} = {2\sqrt{\frac{ɛ}{N_{f}}}{\sum\limits_{n_{f} = 0}^{N_{f}^{\prime} - 1}{{w\left( {t - {2n_{f}T_{f}} - T_{f}} \right)}.}}}} \end{matrix}$ The conditional BER remains the same as equation (21) determined in the original implementation described above.

In another possible implementation, transmitter 4 includes a frame interleaver that permutes the frames of the blocks of symbols to interleave the frames for use in generating the multiple UWB waveforms. For example, a N_(f)×N_(i) frame interleaver may be included in the ST encoder. For example, in the case of ST coding scheme II, the N_(f) repeated versions of s_(a) and s_(b) are fed to the interleaver column-wise and read out row-wise. Choosing the interleaver depth N_(i) to be any even factor of N_(f) can readily modify ST coding scheme II to achieve a diversity order of LN_(i) with two transmit antennas and a MRC-Rake receiver with L fingers. The upper bound of the averaged BER for ST coding scheme II including the frame interleaver is given below.

$\begin{matrix} {{P({error})} \leq \left( {\frac{\beta_{L}}{2N_{i}}\rho} \right)^{{- L}\; N_{i}}} & (33) \end{matrix}$ Compared to equation (21) of ST coding scheme I, frame interleaving for ST coding scheme II achieves a diversity N_(i) times greater. However, frame interleaving also generates a decoding delay of N_(i) frames and a loss in coding gain by a factor of N_(i).

The presented ST coding schemes may also be implemented in a UWB system including more than one receive antenna. Equipping the receiver, with Nr>1 antennas enables receive diversity. Assuming the receive antennas are spaced sufficiently apart so that the channels are mutually independent, receive diversity can be readily exploited with MRC. The upper bound of the averaged BER for ST coding scheme I is given by:

$\begin{matrix} {{P({error})} \leq \left( {\frac{\beta_{L}}{2}\rho} \right)^{{- N_{r}}L}} & (34) \end{matrix}$ and for ST coding scheme II

$\begin{matrix} {{P({error})} \leq \left( {\frac{\beta_{L}}{2N_{i}}\rho} \right)^{{- N_{r}}L\; N_{i}}} & (35) \end{matrix}$ where ST coding scheme II includes a N_(f)×N_(i) frame interleaver. Moreover, because the PAM/PPM UWB transmissions are real by design, the ST coding schemes do not suffer rate loss when more than two transmit antennas are included in a UWB communication system.

FIGS. 6-14 are graphs illustrating results of simulations and comparisons that validate the analyses and designs described above. In all cases, parameters of the channel model are chosen as Γ=33 ns, γ=5 ns, 1/Λ=2 ns, and 1/λ=0.5 ns. The parameter r is chosen to be 0.1225 ns to obtain a pulse width of 0.7 ns. The frame duration is chosen to be T_(f)=100 ns, while the maximum delay spread is T_(g)=99 ns.

FIG. 6 is a graph illustrating results of a comparison of BER performance for single antenna transmission and ST coding schemes I and II with two transmit antennas and one receive antenna. BER vs. SNR curves are plotted in FIG. 6 with the number of fingers, L, of the Rake receiver being 1, 4, and 16. For all values of L, ST coding schemes I and II provide, respectively, twice and four times the diversity order of the single antenna transmission. It can also be seen in FIG. 6 that as the number of Rake receiver fingers L increases, the coding gain is held back.

FIG. 7 is a graph illustrating results of a simulation of the effects of various interleaver depths, N_(i), on BER performance for ST coding scheme II. In this case, ST coding scheme II includes PAM, a Rake receiver with L=1 fingers, two transmit antennas, and one receive antenna. FIG. 7 shows that the diversity order increases with increasing N_(i). In addition, coding gain loss and decoding delay length also increases with increasing N_(i).

FIG. 8 is a graph illustrating results of a comparison of BER performance for ST coding schemes I and II with one receive antenna and with two receive antennas. For both coding schemes the addition of a second receive antenna doubles the diversity gain. However, ST coding scheme I with N_(r)=2 provides the same diversity order, but 3 dB more coding gain than ST coding scheme II with N_(r)=2.

FIGS. 9 and 10 are graphs illustrating results of a simulation of the effects of timing jitter on BER performance for single antenna and multi-antenna transmissions. FIG. 9 shows BER vs. SNR curves for a single transmit and receive antenna system and a two transmit and one receive antenna system employing ST coding scheme II with L=1, 4, and 16 without timing jitter. As can be seen, the diversity gain increases with both L and N_(t). FIG. 10 shows BER vs. SNR curves for the same systems in the presence of timing jitter. Performance degradation is observed for both systems and larger L values are seen to not make much difference in the diversity order. However, the multi-antenna system outperforms the single antenna system for all values of L.

FIGS. 11 and 12 are graphs illustrating results of a comparison of BER performance for single antenna transmission and ST coding scheme I with one receive antenna and two receive antennas. In FIG. 11, ST coding scheme I includes PPM and a Rake receiver with L=1 fingers. The upper curve is obtained with modulation delay Δ=1 ns, which yields an orthogonal PPM. The lower curve is obtained with modulation delay Δ=0.156 ns, which maximizes the correlation. For both delay values, multiple antenna systems provide higher diversity order. The same result can be seen in FIG. 12 for a Rake receiver with L=4 fingers.

FIG. 13 is a graph illustrating results of a comparison of BER performance for ST coding scheme II with one receive antenna and two receive antennas. In this case, ST coding scheme II includes OOK, a special case of PPM with Δ=T_(g)+T_(ω), with coherent reception and a Rake receiver with L=1, 4, and 16 fingers. The diversity order increases with both L and N_(r).

FIG. 14 is a graph illustrating results of a comparison of BER performance for single antenna transmission and ST coding scheme I and II with noncoherent reception. In this case, ST coding schemes I and II include OOK with noncoherent reception and a Rake receiver with L=1, 4, and 16 fingers. The diversity order increases with the coding scheme and with L. However, performance loss is observed when the curves of ST coding scheme II are compared to the coherent reception curves for N_(t)=2 and N_(r)=1 from FIG. 13. The performance loss with noncoherent reception is a tradeoff for the advantage of foregoing channel estimation.

Various embodiments of a multi-antenna UWB communication system including ST coding has been described. As one example, an UWB system including two transmit antennas and one receive antenna has been described. Two exemplary analog ST coding schemes are presented herein in reference to the exemplary two-transmit, one-receive UWB ST-coding system. The ST coding schemes encode analog waveforms within data symbols to allow transmission via multiple antennas. Applying the ST coding schemes to the UWB system enhances spatial and multipath diversity gains without increasing receiver complexity. Furthermore, several possible embodiments of the UWB system and the ST coding schemes have been described.

Nevertheless, various modifications may be made without departing from the techniques described herein. For example, a multi-antenna UWB communication system may be designed using a variety of components and layouts not described herein. Furthermore, any number of transmit and/or receive antennas may be implemented in the UWB system. ST coding schemes other than the examples presented may be used to enable multi-antenna transmission and spatial diversity.

The described techniques can be embodied in a variety of devices that communicate using ultra wideband communication, including wireless personal area networks (WPAN), sensor networks, base stations, mobile phones, laptop computers, handheld computing devices, personal digital assistants (PDA's), and the like. The devices may include a digital signal processor (DSP), field programmable gate array (FPGA), application specific integrated circuit (ASIC) or similar hardware, firmware and/or software for implementing the techniques. If implemented in software, a computer-readable medium may store computer readable instructions, i.e., program code, that can be executed by a processor or DSP to carry out one of more of the techniques described above. For example, the computer-readable medium may comprise random access memory (RAM), read-only memory (ROM), non-volatile random access memory (NVRAM), electrically erasable programmable read-only memory (EEPROM), flash memory, or the like. The computer-readable medium may comprise computer readable instructions that when executed in a wireless communication device, cause the wireless communication device to carry out one or more of the techniques described herein. These and other embodiments are within the scope of the following claims. 

1. A method comprising: processing a stream of information-bearing symbols to form a plurality of symbol blocks, wherein each symbol block comprises more than one of the information bearing symbols; generating multiple ultra-wideband (UWB) waveforms from the symbol blocks, wherein each of the UWB waveforms convey the symbols of their respective symbol blocks as pulses repeated over a plurality of frames; and transmitting the UWB waveforms over different antennas as a space-time coded UWB communication.
 2. The method of claim 1, wherein processing a stream of information-bearing symbols comprises parsing the stream of symbols into blocks of symbol pairs, wherein generating multiple UWB waveforms comprises generating a first UWB waveform to transmit the symbol pairs in a first order and a second UWB waveform to transmit the symbol pairs in a second order opposite from the first order, and wherein transmitting the UWB waveforms comprises simultaneously transmitting the first UWB waveform from a first transmit antenna and the second UWB waveform from a second transmit antenna.
 3. The method of claim 1, wherein processing a stream of information-bearing symbols comprises: parsing the stream into a first block of symbols while maintaining an order of the stream of symbols; and permuting the symbols of the first block to form a second block in which the symbols are in an order different from the order of the stream of symbols.
 4. The method of claim 1, wherein generating multiple UWB waveforms power loading and pulse shaping each of the symbols of the symbol blocks to generate the pulses for transmission repeatedly over the plurality of frames.
 5. The method of claim 1, wherein generating multiple UWB waveforms comprises applying pulse amplitude modulation.
 6. The method of claim 1, wherein generating multiple UWB waveforms comprises applying pulse position modulation.
 7. The method of claim 1, wherein generating multiple UWB waveforms comprises: permuting the frames to interleave the frames; and generating multiple UWB waveforms from the interleaved frames.
 8. The method of claim 1, further comprising: receiving the transmitted UWB waveforms through a wireless communication channel with a plurality of receive antennas; and performing maximum ratio combining (MRC) on the plurality of frames to produce a stream of estimate symbols.
 9. The method of claim 8, wherein receiving the transmitted UWB waveforms comprises: receiving a first UWB waveform of the transmit signals with a receive antenna; receiving a second UWB waveform of the transmit signals with the receive antenna, and wherein performing MRC comprises: performing maximum ratio combining (MRC) on the first UWB waveform to yield a first decision statistic; performing MRC on the second UWB waveform to yield a second decision statistic; combining the first and second decision statistics to create a combined decision statistic; and outputting an estimate symbol based on the combined decision statistic.
 10. The method of claim 8, further comprising separating the received UWB waveforms into even and odd indexed frames at the receive antennas.
 11. A method comprising: processing a stream of information-bearing symbols to form a plurality of symbol blocks, wherein each symbol block comprises one or more of the information bearing symbols; generating multiple ultra-wideband (UWB) waveforms from the symbol blocks, wherein each of the UWB waveforms convey the symbols of their respective symbol blocks as pulses repeated over a plurality of frames; transmitting the UWB waveforms over different antennas as a space-time coded UWB communication, wherein processing a stream of information-bearing symbols comprises duplicating each symbol to form a first symbol block and a second symbol block each comprising the same information bearing symbol, wherein generating multiple UWB waveforms comprises generating a first UWB waveform from the first symbol block and a second UWB waveform from the second symbol block, and wherein transmitting the UWB waveforms comprises simultaneously transmitting the first UWB waveform from a first transmit antenna and the second UWB waveform from a second transmit antenna.
 12. A wireless communication device comprising: a space-time (ST) encoder that processes a stream of information-bearing symbols to form a plurality of ST-encoded symbol blocks, wherein each symbol block comprises more than one of the information bearing symbols; a plurality of pulse shapers that generate multiple ultra-wideband (UWB) waveforms from the symbol blocks, wherein each of the UWB waveforms convey the symbols of their respective symbol blocks as pulses repeated over a plurality of frames; and a plurality of antennas that transmit the UWB waveforms over a wireless communication channel.
 13. The wireless communication device of claim 12, wherein the ST encoder parses the stream of symbols into blocks of symbol pairs and, for each symbol pair, generates a first symbol block that stores the pair of symbols in a first order and a second symbol block that duplicates the pair of symbols and stores the pair of symbols in a second order opposite from the first order.
 14. The wireless communication device of claim 12, further comprising a frame interleaver that permutes the frames to interleave the frames.
 15. The wireless communication device of claim 12, wherein the pulse shapers modulate the pulses for transmission repeatedly over the frames.
 16. The wireless communication device of claim 12, wherein the pulse shapers apply pulse amplitude modulation to the pulses.
 17. The wireless communication device of claim 12, wherein the pulse shapers apply pulse position modulation.
 18. The wireless communication device of claim 12, wherein the wireless communication device comprises one of a base station and a mobile device.
 19. The wireless communication device of claim 12, wherein the ST encoder parses the stream into a first block of symbols while maintaining an order of the stream of symbols, and permutes the symbols of the first block to form a second block in which the symbols are in an order different from the order of the stream of symbols.
 20. A wireless communication device comprising: a space-time (ST) encoder that processes a stream of information-bearing symbols to form a plurality of ST-encoded symbol blocks, wherein each symbol block comprises one or more of the information bearing symbols; a plurality of pulse shapers that generate multiple ultra-wideband (UWB) waveforms from the symbol blocks, wherein each of the UWB waveforms convey the symbols of their respective symbol blocks as pulses repeated over a plurality of frames; a plurality of antennas that transmit the UWB waveforms over a wireless communication channel, wherein the ST encoder duplicates each symbol to form a first symbol block and a second symbol block each comprising the same information bearing symbol, and the plurality of pulse generators generate a first UWB waveform from the first symbol block and a second UWB signal from the second symbol block for simultaneous transmission via the plurality of antennas.
 21. A wireless communication device comprising: a plurality of antennas to receive a plurality of space-time (ST) encoded ultra wideband (UWB) waveforms through a wireless communication channel, each ST encoded UWB waveform having a plurality of information-bearing symbols within a symbol block that are conveyed as pulses repeated over a plurality of frames; and a maximum ratio combining (MRC) unit that processes the ST encoded UWB signals and produces a stream of estimate symbols.
 22. The wireless communication device of claim 21, wherein the received UWB waveforms are separated into even and odd indexed frames at the receive antennas.
 23. The wireless communication device of claim 21, wherein the plurality of antennas comprise a first antenna that receives a first UWB waveform and a second antenna that receives a second UWB waveform, and wherein the MRC unit: performs maximum ratio combining (MRC) on the first UWB waveform to yield a first decision statistic, performs MRC on the second UWB waveform to yield a second decision statistic, combines the first and second decision statistics to create a combined decision statistic, and outputs one of the estimate symbols based on the combined decision statistic.
 24. The wireless communication device of claim 21, wherein the wireless communication device comprises one of a base station and a mobile device.
 25. An ultra-wideband communication system comprising: a transmitter that outputs a plurality of space-time (ST) encoded ultra wideband (UWB) waveforms via a plurality transmit antennas, each ST encoded UWB waveform having a plurality of information-bearing symbols within a symbol block that are conveyed as pulses repeated over a plurality of frames; and a receiver that receives the plurality of ST-encoded UWB waveforms via a wireless communication channel, and performs maximum ratio combining (MRC) on the UWB signals to produce estimate symbols.
 26. The ultra-wideband communication system of claim 25, wherein the receiver comprises a plurality of receive antennas to receive the UWB waveforms.
 27. The ultra-wideband communication system of claim 26, wherein the received UWB waveforms are separated into even and odd indexed frames at the receive antennas. 